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19-0880; Rev 0; 7/07
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
General Description
The MAX16812 is a peak-current-mode LED driver with an integrated 0.2 power MOSFET designed to control the current in a single string of high-brightness LEDs (HBLEDs). The MAX16812 can be used in multiple converter topologies such as buck, boost, or buck-boost. The MAX16812 operates over a 5.5V to 76V wide supply voltage range. The MAX16812 features a low-frequency, wide-range brightness adjustment (100:1), analog and PWM dimming control input, as well as a resistor-programmable EMI suppression circuitry to control the rise and fall times of the internal switching MOSFET. A high-side LED current-sense amplifier and a dimming MOSFET driver are also included, simplifying the design and reducing the total component count. The MAX16812 uses peak-current-mode control, adjustable slope compensation that allows for additional design flexibility. The device has two current regulation loops. The first loop controls the internal switching MOSFET peak current, while the second current regulation loop controls the LED current. Switching frequency can be adjusted from 125kHz to 500kHz. Additional features include adjustable UVLO, soft-start, external enable/disable input, thermal shutdown, a 1.238V 1% accurate buffered reference, and an onchip oscillator. An internal 5.2V linear regulator supplies up to 20mA to power external devices. The MAX16812 is available in a thermally enhanced 5mm x 5mm, 28-pin TQFN-EP package and is specified over the automotive -40C to +125C temperature range.
Features
Integrated 76V, 0.2 (typ) Power MOSFET 5.5V to 76V Wide Input Range Adjustable LED Current with 5% Accuracy Floating Differential LED Current-Sense Amplifier Floating Dimming N-Channel MOSFET Driver PWM LED Dimming with: PWM Control Signal Analog Control Signal Chopped VIN Input Peak-Current-Mode Control 125kHz to 500kHz Adjustable Switching Frequency Adjustable UVLO and Soft-Start Output Overvoltage Protection 5s LED Current Rise/Fall Times During Dimming Minimize EMI Overtemperature and Short-Circuit Protection
MAX16812

Ordering Information
PART MAX16812ATI+ TEMP RANGE -40C to +125C PINPACKAGE PKG CODE
28 TQFN-EP* T2855-8
+Denotes a lead-free package. *EP = Exposed pad.
Simplified Diagram
CH_REG DOUT VOUT
Applications
Automotive Lighting: DRL, Fog Lights Rear Combination Lights Front and Rear Signal Lights Interior Lighting Warning and Emergency Lighting Architectural and Industrial Lighting
CSVIN LV IN
RCS
H_REG
CS+
DGT
DD
HV
LX
COUT RSRC SRC
GT CIN RT RT RTGRM L_REG TGRM CTGRM CS_OUT DIM SGND AGND REFI REF OV COMP FB EN
MAX16812
DRV
SLP CSLP
ROV1 VOUT ROV2 RCOMP1 RCOMP2
CCOMP1
Typical Application Circuit and Pin Configuration appear at end of data sheet.
BUCK-BOOST CONFIGURATION
________________________________________________________________ Maxim Integrated Products
1
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
ABSOLUTE MAXIMUM RATINGS
(All voltages are referenced to AGND, unless otherwise noted.) SGND ....................................................................-0.3V to +0.3V IN, EN, LX, DIM ......................................................-0.3V to +80V L_REG, GT, DRV ......................................................-0.3V to +6V RT, REF, REFI, CS_OUT, FB, COMP, SRC, SLP, TGRM, OV ....................................................-0.3V to +6V LV, HV, CS-, CS+, DGT, DD, H_REG ....................-0.3V to +80V CS+, DGT, H_REG to LV ........................................-0.3V to +12V CS- to LV ...............................................................-0.3V to +0.3V CS+ to CS- .............................................................-0.3V to +12V *As per JEDEC51 standard (multilayer board).
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
DD to LV ....................................................................-1V to +80V Maximum Current into Any Pin (except LX, SRC) ............20mA Maximum Current into LX and SRC.......................................+2A Continuous Power Dissipation (TA = +70C) 28-Pin TQFN 5mm x 5mm (derate 34.65mW/C* above +70C) .........................2759mW Operating Temperature Range .........................-40C to +125C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
ELECTRICAL CHARACTERISTICS
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, CREF = 47nF, VTGRM = 0V, RSRC = 0.2, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER Input Voltage Range Quiescent Supply Shutdown Supply Current Internal MOSFET On-Resistance Output Current Accuracy Peak Switch Current Limit Hiccup Switch Current Switch Leakage Current UNDERVOLTAGE LOCKOUT IN Undervoltage Lockout UVLO Hysteresis EN Threshold Voltage EN Hysteresis REFERENCE (REF) AND LOW-SIDE LINEAR REGULATOR (L_REG) Startup Response Time Reference Voltage Reference Soft-Start Charging Current L_REG Supply Voltage L_REG Load Regulation L_REG Dropout Voltage tPOR VREF IREF_SLEW VIN or VEN rising IREF = 10A VREF = 0V VIN = 7.5V, IL_REG = 1mA IL_REG = 20mA IL_REG = 25mA 400 1.190 25 4.9 50 1.238 40 5.2 1.288 60 5.5 20 s V A V mV VEN_THUP VEN rising 1.2 UVLO VIN rising 4.6 4.9 100 1.38 100 1.6 5.3 V mV V mV ILXLEAK VEN = 0V, VLX = 76V, VGT = 0V SYMBOL VIN IQ ISHDN RDSON ILED ILXLIM VTGRM = 1V, VDIM = 0V VEN 300mV ILX = 1A, VIN > 10V, VGT = VDRV = 5V ILED = 350mA, RCS = 1 -5 2.6 3.1 6 1 10 CONDITIONS MIN 5.5 0.3 20 0.2 TYP MAX 76.0 2.5 45 0.4 +5 3.6 UNITS V mA A % A A A
2
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, CREF = 47nF, VTGRM = 0V, RSRC = 0.2, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER PWM COMPARATOR COMP Input Leakage Current SRC Input Leakage Current Comparator Offset Voltage Input Voltage Range Propagation Delay ERROR AMPLIFIER FB Input Current REFI Input Current Error-Amplifier Offset Voltage Input Common-Mode Range Source Current Sink Current COMP Clamp Voltage DC Gain Unity-Gain Bandwidth VCOMP ICOMP VOS VFB = 1V, VREFI = 1.2V VFB = 1V, VREFI = 1V VFB = VCOMP = 1.2V VFB = (VCOMP - 0.9V) (VREFI - VFB) 0.5V (VFB - VREFI) 0.5V VREF = 1.2V, VFB = 0V -100 -100 -23 0 300 80 1.20 72 0.8 2.56 +100 +100 +23 1.5 nA nA mV V A A V dB MHz ILKCOMP ILKSRC VOS(EA) VSRC tPD VCOMP = 1V, VSRC = 0.5V, VTGRM = 1V, VDIM = 0.5V VCOMP = 0V, VSRC = 0.5V, VTGRM = 0V, VDIM = 0.5V (VCOMP - VSRC) = VOS VCOMP = VSRC + 860mV 50mV overdrive 0 100 -0.10 -5 860 1.23 +0.10 +5 A A mV V ns SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX16812
ELECTRICAL CHARACTERISTICS
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, CREF = 47nF, VTGRM = 0V, RSRC = 0.2, RCS = 1, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER H_REG Input-Voltage Threshold H_REG Supply Voltage H_REG Load Regulation Dropout Voltage CS- Input Bias Current CS+ Input Bias Current Input Voltage Range Minimum Output Current Output Voltage Range DC Voltage Gain Unity-Gain Bandwidth Maximum REFI Input Voltage VREFI ICS_OUT VCS_OUT ICSICS+ SYMBOL CONDITIONS VH_REG is rising IH_REG = 0 IH_REG = 0 to 3mA IH_REG = 5mA VCS- = VLV, (VCS+ - VCS-) = -0.1V VCS- = VLV, (VCS+ - VCS-) = 0.1V VCS- = VLV Sinking Sourcing -1 0 25 400 0 4 0.8 1.0 1.5 820 500 +1 0.25 MIN 3.60 4.75 TYP 3.887 5 MAX 4.20 5.40 80 UNITS V V mV A A V A V V/V MHz V
HIGH-SIDE UNDERVOLTAGE LOCKOUT AND LINEAR REGULATOR (H_REG) ((VHV - VLV) = 21V)
HIGH-SIDE CURRENT-SENSE AMPLIFIERS (VHV - VLV) = 21V
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, CREF = 47nF, VTGRM = 0V, RSRC = 0.2, RCS = 1, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
HIGH-SIDE DIMMING LINEAR REGULATOR ((VHV - VLV) = 21V) VLV = VCS-, (VCS+ - VCS-) = 0.3V, (VDD - VLV) = 1V, VDIM = 1V, VTGRM = 0V, VDGT = 1V, VREFI = 1.0V, sinking Minimum Output Current IDGT VLV = VCS-, (VCS+ - VCS-) = 0.2V, (VDD - VLV) = 1V, VTGRM = 0V, VDGT = 3V, VREFI = 1.0V, VDIM = 1V, sourcing Output Voltage Range DC Gain DD Input Bias Current DD Input Low Threshold DIMMING ((VHV - VLV) = 21V) DIM Input Bias Current TGRM Input High Threshold TGRM Reset High-to-TGRM Low Pulse Width TGRM Reset Switch RDS(ON) Dimming Rise and Fall LED Current Times OVERVOLTAGE PROTECTION (OV) OV Input High Threshold OV Input Threshold Hysteresis OV Input Bias Current INTERNAL OSCILLATOR CLOCK Internal Clock Frequency fOSC RT = 2M to AGND RT = 50k to AGND VSLP = 0V DRV sinking 20mA DRV sourcing 20mA VGT = 0 to 5V VTH Qg VLX = 50V 3 10 -1 2.5 8 470 105 525 125 150 30 45 +1 570 155 kHz IOV VOV = 1.1V -1 VOV rising 1.180 1.230 14 +1 1.292 V mV A VTGRM = 1.3V 5 IDIM VDIM = 1.1V -1 1.18 1.23 1 20 +1 1.27 A V s s IDD CDGT = 1nF to LV (VDD - VCS-) = 0.5V VTGRM = 0V, VDIM = 1V, VREFI = 1.2V, (VDGT - VLV) > 1.5V, VDD falling -3 0.25 0.50 0.2 60 +3 0.75 1.2 1.2 mA
5.0
V dB A V
SLOPE COMPENSATION INPUT (SLP) SLP Input Current LOW-SIDE GATE DRIVE (DRV) DRV Output Low Impedance DRV Output High Impedance INTERNAL POWER MOSFET GT Input Leakage Current Internal MOSFET Gate-to-Source Threshold Voltage Internal MOSFET Gate Charge A V nC RDRV_LO RDRV_HI ISLP A
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
Typical Operating Characteristics
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, VTGRM = 0V, TA = +25C, unless otherwise noted.)
SWITCH CURRENT LIMIT vs. TEMPERATURE
MAX16812 toc02
MAX16812
RDS(ON) vs. ILX
MAX16812 toc01
RDS(ON) vs. VGT
2.0 1.8 1.6 1.4 RDS(ON) () 1.2 1.0 0.8 0.6 0.4 0.2 0 TA = +25C 3.300 3.250 SWITCH CURRENT LIMIT (A) 3.200 3.150 3.100 3.050 3.000 2.950 2.900 2.2 2.8 3.4 4.0 4.6 VGT (V) 5.2 5.8 6.4 7.0
0.40 0.35 0.30 RDS(ON) () 0.25 0.20 0.15 0.10 0.05 0 1.0 1.5 2.0 ILX (A) 2.5 TA = -40C TA = +25C TA = +125C
3.0
-40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
SHUTDOWN CURRENT vs. TEMPERATURE
MAX16812 toc04
VREF vs. TEMPERATURE
MAX16812 toc05
IN UVLO THRESHOLD vs. TEMPERATURE
VIN RISING
MAX16812 toc06
30 25 SHUTDOWN CURRENT (A) 20
1.25
5.20
15 10
1.23
IN UVLO THRESHOLD (V)
1.24 VREF (V)
5.15
5.10
1.22 5 IREF = 10A 0 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C) 1.21 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
5.05
5.00 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
IN UVLO THRESHOLD vs. TEMPERATURE
MAX16812 toc07
EN UVLO THRESHOLD vs. TEMPERATURE
1.45 1.40 1.35 EN UVLO (V) 1.30 1.25 1.20 1.15 1.10 1.05 1.00 VEN RISING
MAX16812 toc08
5.10 5.09 5.08 5.07 IN UVLO (V) 5.06 5.05 5.04 5.03 5.02 5.01 5.00 VIN FALLING
1.50
-40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
-40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
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MAX16812 toc03
0.45
5
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
Typical Operating Characteristics (continued)
(VIN = VEN = 12V, CL_REG = 3.3F, CH_REG = 1F, VTGRM = 0V, TA = +25C, unless otherwise noted.)
EN UVLO THRESHOLD vs. TEMPERATURE
MAX16812 toc09
VL_REG vs. IL_REG
MAX16812 toc10
OSCILLATOR FREQUENCY vs. TEMPERATURE
RT = 2M OSCILLATOR FREQUENCY (kHz) 500 400 RT = 180k 300 200 RT = 50k 100 VIN = 7.5V 0 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
MAX16812 toc11 MAX16812 toc13
1.50 1.45 1.40 1.35 EN UVLO (V) VEN FALLING
5.5 5.4 5.3 5.2 VL_REG (V) 5.1 5.0 4.9 4.8 4.7 4.6 4.5 TA = -40C TA = +25C TA = +125C
600
1.30 1.25 1.20 1.15 1.10 1.05 1.00 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
0
2
4
6
8
10 12 14 16 18 20
IL_REG (mA)
OSCILLATOR FREQUENCY vs. RT
MAX16812 toc12
VH_REG THRESHOLD vs. TEMPERATURE
4.2 4.1 VH_REG THRESHOLD (V) 4.0 3.9 3.8 3.7 3.6
600 OSCILLATOR FREQUENCY (kHz) 500 400 300 200 100
3.5 0 0.01 0.1 RT (M) 1 10 3.4 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
VH_REG vs. IH_REG
MAX16812 toc14
VH_REG vs. TEMPERATURE
5.1 5.0 4.9 VH_REG (V) 4.8 4.7 4.6 4.5 (VHV - VLV) = 21V ILOAD = 3mA
MAX16812 toc15
5.00 4.95 4.90 4.85 VH_REG (V) 4.80 4.75 4.70 4.65 4.60 4.55 4.50 0 0.5 1.0 1.5 IH_REG (mA) 2.0 2.5 VH_REG IS MEASURED WITH RESPECT TO VLV (VHV - VLV) = 6V VIN = 12V
5.2
4.4 4.3
3.0
4.2 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (C)
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
Pin Description
PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20, 21 22, 23 24 25 26 27 NAME FB COMP REFI REF CS_OUT AGND EN IN L_REG SGND DD DGT CS+ CSLV H_REG HV DRV GT LX SRC SLP TGRM DIM RT Low-Side Error Amplifier's Inverting Input Low-Side Error Amplifier's Output. Connect a compensation network from COMP to FB for stable operation. Reference Input. VREFI provides the reference voltage for the high-side current-sense amplifier to set the LED current. +1.23V Reference Output. Connect an appropriate soft-start capacitor from REF to AGND. High-Side Current-Sense Amplifier Output. VCS_OUT is proportional to the current through RCS. Analog Ground Enable Input/Undervoltage Lockout. Connect EN to IN through a resistive voltage-divider to program the UVLO threshold. Connect EN directly to IN to set up the device for 5V internal threshold. Apply a logiclevel input to EN to enable/disable the device. Positive Power-Supply Input. Bypass with a 1F ceramic capacitor to AGND. 5V Low-Side Regulator Output. Bypass with a 3.3F ceramic capacitor to AGND. Signal Ground MOSFET's Drain Voltage-Sense Input. Connect DD to the drain of the external dimming MOSFET. External Dimming MOSFET's Gate Drive High-Side Current-Sense Amplifier's Positive Input. Connect RCS between CS+ and CS-. CS+ is referenced to LV. High-Side Current-Sense Amplifier's Negative Input. Connect RCS between CS- and CS+. CS- is referenced to LV. High-Side Reference Voltage Input. A DC voltage at LV sets the lowest reference point for the high-side current-sense and dimming MOSFET control circuitry. High-Side Regulator Output. H_REG provides a regulated supply for high-side circuitry. Bypass with a 1F ceramic capacitor to LV. High-Side Positive Supply Voltage Input. HV provides power for dimming and LED current-sense circuitry. HV is referenced to LV. Internal MOSFET Gate Driver Output. Connect to a resistor between DRV and GT to set the rise and fall times at LX. Internal MOSFET GATE. Connect a resistor between GT and DRV to set the rise and fall times at LX. Internal MOSFET Drain Internal Power MOSFET Source Slope Compensation Setting. Connect an appropriate external capacitor from SLP to AGND to generate a ramp signal for stable operation. Dimming Comparator's Reference/Ramp Generator Dimming Control Input Resistor-Programmable Internal Oscillator Setting. Connect a resistor from RT to AGND to set the internal oscillator frequency. Overvoltage Protection Input. Connect OV to HI through a resistive voltage-divider to AGND to set the overvoltage limit for the load. When the voltage at OV exceeds the 1.238V (typ) threshold, the gate drive (DRV) for the switching MOSFET is disabled. Once VOV goes below 1.238V by 14mV, the switching MOSFET turns on again. Exposed Pad. Connect EP to a large-area ground plane for effective power dissipation. Do not use as the IC ground connection. FUNCTION
MAX16812
28
OV
--
EP
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
DD DS HV DRMP H_REG LDOH POR 3.88V 1.2X DIM CS+ RAMP 1.1X CMP IHI CSA CSLX LX 1X LV tD = 200ns IN PREG BG VREF VDD LDOL L_REG EN UVLO/ POR S R HICCUP REF 1X EN 0.6V RT DIM CMP OSC DIM SIGNAL LOGIC CONTROL ILIM Q G1 1.2V SGND DRV VREFI = 1.2V VRAMP = 0.3V SRC 2.5V GT SRC ADIM DGT CMP 0.5V
REF
LATCH
VBE
1.238V CMP TGRM
PWM
X0.2
SLP
MAX16812
2s PULSE LOW TO DISCHARGE OV OVP 1.238V SGND AGND X1 ERROR AMPLIFIER AND DIMMING S/H
COMP FB CS_OUT REFI
Figure 1. Functional Diagram
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
Detailed Description
The MAX16812 is a current-mode PWM LED driver with an integrated 0.2 power MOSFET for use in driving HBLEDs. By using two current regulation loops, 5% LED current accuracy is achieved. One current regulation loop controls the internal MOSFET peak current through a sense resistor (RSRC) from SRC to ground, while the other current regulation loop controls the average LED current in a single LED string through another sense resistor (RCS) in series with the LEDs. The MAX16812 includes a cycle-by-cycle current limit that turns off the gate drive to the internal MOSFET during an overcurrent condition. The MAX16812 features a programmable oscillator that simplifies and optimizes the design of magnetics. The MAX16812 is well suited for inputs from 5.5V to 76V. An external resistor in series with the internal MOSFET gate can control the rise and fall times on the drain of the internal switching MOSFET, therefore minimizing EMI problems. The MAX16812 high-frequency, current-mode PWM HBLED driver integrates all the necessary building blocks for driving a series LED string in an adjustable constant current mode with PWM dimming. Currentmode control with leading-edge blanking simplifies control-loop design, and an external adjustable slopecompensation control stabilizes the inner current-mode loop when operating at duty cycles above 50%. An input undervoltage lockout (UVLO) programs the input supply startup voltage. An external voltagedivider on EN programs the supply startup voltage. If EN is directly connected to the input, the UVLO is set at 5V. A single external resistor from RT to AGND programs the switching frequency from 125kHz to 500kHz. Wide contrast (100:1) PWM dimming can be achieved with the MAX16812. A DC input on DIM controls the dimming duty cycle. The dimming frequency is set by the sawtooth ramp frequency on TGRM (see the PWM Dimming section). In addition, PWM dimming can be achieved by applying a PWM signal to DIM with TGRM set to a DC voltage less than 1.238V. A floating highvoltage driver drives an external n-channel MOSFET in series with the LED string. REFI allows analog dimming of the LED current, further increasing the effective dimming range over PWM alone. The MAX16812 has a 5s preprogrammed LED current rise and fall time. A nonlatching overvoltage protection limits the voltage on the internal switching MOSFET under open-circuit conditions in the LED string. The internal thermal shutdown circuit protects the device if the junction temperature should exceed +165C.
Current-Mode Control
The MAX16812 offers a current-mode control operation feature with leading-edge blanking that blanks the sensed current signal applied to the input of the PWM current-mode comparator. In addition, a current-limit comparator monitors the same signal at all times and provides cycle-by-cycle current limit. An additional hiccup comparator limits the absolute peak current to two times the cycle-by-cycle current limit. The leading-edge blanking of the current-sense signal prevents noise at the PWM comparator input from prematurely terminating the on-cycle. The switch current-sense signal contains a leading-edge spike that results from the MOSFET gate-charge current, and the capacitive and diode reverse-recovery current of the power circuit. The MAX16812's capacitor-adjustable slope-compensation feature allows for easy stabilization of the inner switching MOSFET current-mode loop. Upon triggering the hiccup current limit, the soft-start capacitor on REF is discharged and the gate drive to DRV is disabled. Once the inductor current falls below the hiccup current limit, the soft-start capacitor is released and it begins to charge after 10s.
MAX16812
Slope Compensation
The MAX16812 uses an internal ramp generator for slope compensation. The internal ramp signal resets at the beginning of each cycle and slews at the rate programmed by the external capacitor connected at SLP and an internal ISLP current source of 150A. An internal attenuator attenuates the actual slope compensation signal by a factor of 0.2. Adjust the MAX16812 slew-rate capacitor by using the following equation: I CSLOPE = 0.2 x SLP SR where ISLP is the charging current in mA and CSLOPE is the slope compensation capacitance on the SLP in F, and SR is the designed slope in mV/s. When using the MAX16812 for internal switching MOSFET duty cycles greater than 50%, the following conditions must be met to avoid current-loop subharmonic oscillations. SR 0.5 x RSRC x VIND _ OFF L mV / s
where RSRC is in m, VIND_OFF is in volts, and L is in H. L is the inductor connected to the LX pin of the internal switching MOSFET and VIND_OFF is the voltage across the inductor during the off-time of the internal MOSFET.
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
Undervoltage Lockout
The MAX16812 features an adjustable UVLO through the enable input (EN). Connect EN directly to IN to use the 5V default UVLO. Connect EN to IN through a resistive divider to ground to set the UVLO threshold. The MAX16812 is enabled when VEN exceeds the 1.38V (typ) threshold. Calculate the EN UVLO resistor-divider values as follows (see Figure 2): VEN RUV1 = RUV2 x VUVLO - VEN where RUV1 is in the 20k range, VEN is the 1.38V (typ) EN threshold voltage, and VUVLO is the desired inputvoltage UVLO threshold in volts. Due to the 100mV hysteresis of the UVLO threshold, capacitor C EN is required to prevent chattering at the UVLO threshold due to line impedance drops at power-up and during dimming. If the undervoltage setting is very close to the required minimum operating voltage, there can be jumps in the voltage at IN while dimming. CEN should be large enough to limit the ripple on EN to less than 100mV (EN hysteresis) under these conditions so that it does not turn on and off due to the ripple on IN. minimizing output-voltage overshoot. While the part is in UVLO, CREF is discharged (Figure 3). Upon coming out of UVLO, an internal current source starts charging CREF during the soft-start cycle. Use the following equation to calculate total soft-start time: t ST = CREF x 1.238 IREF
where IREF is 40A, CREF is in F, and tST is in seconds. Operation begins when REF ramps above 0.6V. Once the soft-start is complete, REF is regulated to 1.238V, the internal voltage reference.
Low-Side Internal Switching MOSFET Driver Supply (L_REG)
L_REG is the regulated (5.2V) internal supply voltage capable of delivering 20mA. L_REG provides power to the gate drive of the internal switching power MOSFET. V L_REG is referenced to AGND. Connect a 3.3F ceramic capacitor from L_REG to AGND.
High-Side Regulator (H_REG)
H_REG is a low-dropout linear regulator referenced to LV. H_REG provides the gate drive for the external n-channel dimming MOSFET and also powers up the MAX16812's LED current-sense circuitry. Bypass H_REG to LV with a 1F ceramic capacitor.
Soft-Start
The soft-start feature of the MAX16812 allows the LED string current to ramp up in a controlled manner, thus
VIN
VIN
IN RUV2
IN
MAX16812
EN REF
MAX16812
CEN
RUV1 AGND
CREF AGND
Figure 2. UVLO Threshold Setting
Figure 3. Soft-Start Setting
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Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
High-Side Current-Sense Output (CS_OUT)
A high-side transconductance amplifier converts the voltage across the LED current-sense resistor (RCS) into an internal current output. This current flows through an internal resistor connected to AGND. The voltage gain for the LED current-sense signal is 4. The amplified signal is then buffered and connected through an internal switch to CS_OUT. is low, COMP is disconnected from the output of the error amplifier and CS_OUT is simultaneously disconnected from the buffered LED current-sense output signal (Figure 5). When the internal dimming signal is high, the output of the op amp is connected to COMP and CS_OUT is connected to the buffered LED currentsense signal at the same time (Figure 4). This enables the compensation capacitor to hold the charge when the DIM signal has turned off the internal switching MOSFET gate drive. To maintain the charge on the compensation capacitors CCOMP1 and CCOMP2, the capacitors should be of the low-leakage ceramic type. When the internal dimming signal is enabled, the voltage on the compensation capacitor forces the converter into steady state almost instantaneously. The voltage on COMP is subtracted from the internal slope compensation signal and is then connected to one of the inputs of the PWM comparator. The PWM comparator input is of the CMOS type with very low bias currents.
CCOMP2 STATE A RCOMP2 CCOMP1
MAX16812
Internal Error Amplifier
The MAX16812 includes a built-in voltage-error amplifier, which can be used to close the feedback loop. The internal LED current-sense output signal is buffered internally and then connected to CS_OUT through an internal switch. CS_OUT is connected to the inverting input (FB) pin of the error amplifier through a resistor. See Figures 4 and 5. The reference voltage for the output current is connected to REFI, the noninverting input of the error amplifier. When the internal dimming signal
OUT X1
RCOMP1 EA REFI COMP
Figure 4. Internal Error Amplifier Connection (Dimming Signal High)
CCOMP2 STATE B RCOMP2 CCOMP1
OUT X1
RCOMP1 EA REFI COMP
Figure 5. Internal Error Amplifier Connections (Dimming Signal Low)
______________________________________________________________________________________ 11
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
Analog Dimming
The MAX16812 offers analog dimming of the LED current by allowing the application of an external voltage at REFI. The output current is proportional to the voltage at REFI. Use a potentiometer from REF or directly apply an external voltage source at REFI. voltage produced by this current (through the currentsense resistor) exceeds the current-limit (ILIM) comparator threshold, the MOSFET driver (DRV) quickly terminates the current on-cycle. The 200ns leadingedge blanking circuit suppresses the leading-edge spike on the current-sense waveform from appearing at the current-limit comparator. There is also a hiccup comparator (HICCUP) that limits the peak current in the internal switch set at twice the peak limit setting.
PWM Comparator
The PWM comparator uses the instantaneous switch current, the error-amplifier output, and the slope compensation to determine when the gate drive DRV to the internal n-channel switching MOSFET turns off. In normal operation, gate drive DRV to the n-channel MOSFET turns off when: ISW x RSRC VCOMP - VOFFSET - VSCOMP where ISW is the current through the internal n-channel switching MOSFET, RSRC is the switch current-sense resistor, VCOMP is the output voltage of the internal amplifier, VOFFSET is the internal DC offset, which is a VBE drop, and VSCOMP is the ramp function that starts at zero and slews at the programmed slew rate (SR).
Internal n-Channel Switching MOSFET Driver (DRV)
L_REG provides power for the DRV output. Connect a resistor from DRV to gate GT of the internal switching MOSFET to control the switching MOSFET rise and fall times, if necessary.
External Dimming MOSFET Gate Drive (DGT)
DGT is the gate drive to the external dimming MOSFET referenced to LV. H_REG provides the power to the gate drive.
Internal Switching MOSFET Current Limit
The current-sense resistor (RSRC), connected between the source of the internal MOSFET and ground, sets the current limit. The SRC input has a voltage trip level (VSRC) of 600mV for the cycle-by-cycle current limit. Use the following equation to calculate the value of RSRC: RSRC = VSRC ILXLIM
Overvoltage Protection
The overvoltage protection (OVP) comparator compares the voltage at OV with a 1.238V (typ) internal reference. When the voltage at OV exceeds the internal reference, the OVP comparator terminates PWM switching and no further energy is transferred to the load. Connect OV to HV through a resistive voltage-divider to ground to set the overvoltage threshold at the output.
Setting the Overvoltage Threshold
Connect OV to HV or to the high-side of the LEDs through a resistive voltage-divider to set the overvoltage threshold at the output (Figure 6).
where ILXLIM is the peak current that flows through the switching MOSFET at full load and low line. When the
VLED+ HV ROV1 OV ROV2 ROV2 VLED+
MAX16812
ROV1 OV
MAX16812
AGND
AGND
Figure 6. OVP Setting
12
______________________________________________________________________________________
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
The overvoltage protection (OVP) comparator compares the voltage at OV with a 1.238V (typ) internal reference. Use the following equation to calculate resistor values: VOV_LIM - VOV ROV1 = ROV2 x VOV where VOV is the 1.238V OV threshold. Choose ROV1 and ROV2 to be reasonably high-value resistors to prevent the discharge of filter capacitors. This prevents degraded performance during dimming.
RDIM1
MAX16812
REF
L_REG RTGRM
MAX16812
DIM TGRM
RDIM2
CTGRM AGND
Internal Oscillator Switching Frequency
The oscillator switching frequency is programmed by a resistor connected from RT to AGND. To program the oscillator frequency above 125kHz, choose the appropriate resistor RT from the curves shown in the Oscillator Frequency vs. R T graph in the Typical Operating Characteristics section.
Figure 7. PWM Dimming from REF
PWM Dimming
PWM dimming can be achieved by driving DIM with an analog voltage less than VREF. See Figure 7. An external resistor on TGRM from L_REG in conjunction with the ramp capacitor, CTGRM, from TGRM to AGND creates a sawtooth ramp that is compared with the DC voltage on DIM. The output of the comparator is a pulsating dimming signal. The frequency f RAMP of the sawtooth signal on TGRM is given by: fRAMP 3.67 CTGRM x RTGRM
Use the following formula to calculate the voltage VDIM, necessary for a given output duty cycle, D: VDIM = D x 1.238V where VDIM is the DC voltage applied to DIM in volts. The DC voltage for DIM can also be created by connecting DIM to REF through a resistive voltage-divider. Using the required dimming input voltage, VDIM, calculate the resistor values for the divider string using the following equation: RDIM2 = [VDIM / (VREF - VDIM)] x RDIM1 where VREF is the voltage on REF.
PWM dimming can also be achieved by connecting TGRM to a DC voltage less than VREF and applying the PWM signal at DIM. The moment the internal dimming signal goes low, gate drive DRV to the internal switching MOSFET is turned off. The error amplifier goes to state B (see the Internal Error Amplifier section and Figures 4 and 5). The peak current in the inductor prior to disabling DRV is ILX. Gate drive DGT to the external dimming MOSFET is held high. Then after a switchover period, gate voltage V DGT on the external dimming MOSFET is linearly controlled to reduce the LED current to 0. The fall time of the LED current is controlled by an internal timing circuit to 5s for the MAX16812. During this period, the gate (DRV) to the internal switching MOSFET is enabled. After the fall time, the gate drive to the external dimming MOSFET is turned off and the gate drive to the internal switching MOSFET is still held high after the switchover period. The peak current in the inductor is controlled at ILX. Then after a time period of 20s, the gate drive is disabled. The scope shots in Figures 8-11 show the dimming waveforms.
______________________________________________________________________________________
13
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
MAX16812 fig08 MAX16812 fig10
VOUT ILED
10V/div 100mA/div 0A, 0V
VOUT
10V/div 100mA/div
ILED
0A, 0V
2V/div VDRV 0V VDRV
2V/div 0V
10s/div
10s/div
Figure 8. LED Current, Output Voltage, and DRV Waveforms when DIM Signal Goes Low
Figure 10. LED Current, Output Voltage, and DRV Waveforms when DIM Signal Goes High
MAX16812 fig09
MAX16812 fig11
ILED ILED 100mA/div 100mA/div VDIM 5V/div 0A, 0V VDRV 2V/div 0V 0V
VDIM 5V/div
0A, 0V
VDRV 2V/div
10s/div
10s/div
Figure 9. LED Current, DIM Signal, and DRV Waveforms when DIM Signal Goes Low
Figure 11. LED Current, DIM Signal, and DRV Waveforms when DIM Signal Goes High
When the DIM signal goes high, the LED current is gradually increased to the programmed value. The rise time of the LED current is controlled to 5s for the MAX16812 by controlling the voltage on DGT. After the rise time, an internal sensing circuit monitors the voltage across the drain to the source of the external dimming MOSFET. The LED current is now controlled at the programmed value by a linear current regulating circuit. Once the voltage across the drain to source of the dimming MOSFET drops below 0.5V, the reference for the linear current regulating circuit is increased to 1.1 times the programmed value. The gate drive (DRV) to the internal switching MOSFET is enabled and the error amplifier is returned to state A (see the Internal Error Amplifier section and Figures 4 and 5).
Fault Protection
The MAX16812 features built-in overvoltage protection and thermal shutdown. Connect a resistive voltagedivider between HV, OV, and AGND to program the overvoltage protection. In the case of a short circuit across the LED string, the temperature of the external dimming MOSFET could exceed the maximum allowable junction temperature. This is due to excess power dissipation in the MOSFET. Use the fault protection circuit shown in Figure 12 to protect the external dimming MOSFET. Internal thermal shutdown in the MAX16812 safely turns off the IC when the junction temperature exceeds +165C.
14
______________________________________________________________________________________
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
VIN
100k GND TOVER GND 5.1V ZENER TO EN PIN OF MAX16812
MAX6501
TO L_REG PIN OF MAX16812
VCC
4.7F
Figure 12. Dimming MOSFET Protection
Inductor Selection
The minimum required inductance is a function of the operating frequency, the input-to-output voltage differential and the peak-to-peak inductor current (I L ). Higher IL allows for a lower inductor value while a lower I L requires a higher inductor value. A lower inductor value minimizes size and cost, improves largesignal transient response, but reduces efficiency due to higher peak currents and higher peak-to-peak output ripple voltage for the same output capacitor. On the other hand, higher inductance increases efficiency by reducing the ripple current, I L. However, resistive losses due to the extra turns can exceed the benefit gained from lower ripple current levels, especially when the inductance is increased without allowing for larger inductor dimensions. A good compromise is to choose IL equal to 30% of the full load current. The inductor saturating current specification is also important to avoid runaway current during output overload and continuous short-circuit conditions. Buck Configuration: In a buck configuration (Figure 13), the average inductor current does not vary with the input. The worst-case peak current occurs at the highest input voltage. In this case, the inductance, L, for continuous conduction mode is given by: L= VOUT x (VINMAX - VOUT ) VINMAX x fSW x IL
where VINMAX is the maximum input voltage, fSW is the switching frequency, and VOUT is the output voltage. Boost Configuration: In the boost converter, the average inductor current varies with the input voltage and the maximum average current occurs at the lowest input voltage. For the boost converter, the average inductor current is equal to the input current. In this case, the inductance, L, is calculated as: L= VINMIN x (VOUT - VINMIN ) VOUT x fSW x IL
where VINMIN is the minimum input voltage, VOUT is the output voltage, and fSW is the switching frequency. See Figure 14. Buck-Boost Configuration: In a buck-boost converter (see the Typical Application Circuit), the average inductor current is equal to the sum of the input current and the LED current. In this case, the inductance, L, is: L= VOUT x VINMIN
(VOUT + VINMIN ) x fSW x IL
where VINMIN is the minimum input voltage, VOUT is the output voltage, and fSW is the switching frequency.
______________________________________________________________________________________
15
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
COUT VIN DOUT CH_REG RCS
CIN
IN H_REG EN
HV
LX
LV
DD
DGT CS-
CS+ SRC RSRC
RRT RT CL_REG L_REG RTGRM TGRM CTGRM DIM OV VOUT ROV1 ROV2 RREF2 CCOMP2 SGND AGND REF REFI CS_OUT COMP FB GT
MAX16812
DRV CSLP SLP
RG
CREF
RCOMP1 RREF1 RCOMP2
CCOMP1
Figure 13. Buck Configuration
CH_REG
RCS
DOUT
VOUT
VIN CSVIN LV IN CIN1 RRT CL_REG L_REG RTGRM TGRM CTGRM DIM OV VOUT ROV1 ROV2 RREF2 CCOMP2 SGND AGND REF REFI CS_OUT COMP FB EN RT DRV CS+ DGT DD H_REG HV LX SRC GT RG RSRC COUT
MAX16812
SLP CSLP
CREF
RCOMP1 RREF1 RCOMP2
CCOMP1
Figure 14. Boost Configuration
16 ______________________________________________________________________________________
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
L1 L2 CH_REG DOUT RCS DD DGT CS+ CSVIN H_REG HV LX RSRC COUT CS VOUT
VIN
LV IN CIN1 RT CL_REG L_REG RTGRM CTGRM EN RT
SRC GT RG
MAX16812
DRV SLP CSLP CS_OUT
SGND
AGND
TGRM DIM OV
REFI
REF
COMP FB
VOUT ROV1 ROV2 RREF2 RREF1 RCOMP1 CCOMP2 RCOMP2 CCOMP1
Figure 15. SEPIC Configuration
Output Capacitor
The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and the bulk capacitance of the output capacitor contribute to the output ripple. In most of the applications, the output ESR and ESL effects can be dramatically reduced by using low-ESR ceramic capacitors. To reduce the ESL effects, connect multiple ceramic capacitors in parallel to achieve the required capacitance. In a buck configuration, the output capacitance, COUT, is calculated using the following equation: COUT (VINMAX - VOUT ) x VOUT VR x 2 x L x VINMAX x fSW 2
In a buck-boost configuration, the output capacitance, COUT is: COUT 2 x VOUT x IOUT VR x (VOUT + VINMIN ) x fSW
where VOUT is the voltage across the load and IOUT is the output current.
Input Capacitor
An input capacitor connected between IN and ground must be used when configuring the MAX16812 as a buck converter. Use a low-ESR input capacitor that can handle the maximum input RMS ripple current. Calculate the maximum RMS ripple using the following equation: IIN(RMS) = IOUT x VOUT x (VINMIN - VOUT ) VINMIN
where VR is the maximum allowable output ripple. In a boost configuration, the output capacitance, COUT, is calculated as: COUT (VOUT - VINMIN ) x 2 x IOUT VR x VOUT x fSW
where COUT is the output capacitor.
When using the MAX16812 in a boost or buck-boost configuration, the input capacitor's RMS current is low and the input capacitance can be small. However, an additional electrolytic capacitor may be required to prevent oscillations due to line impedances.
17
______________________________________________________________________________________
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
Layout Recommendations
Typically, there are two sources of noise emission in a switching power supply: high di/dt loops and high dv/dt surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the drain of the internal MOSFET connected to the LX pin presents a dv/dt source. Keep all PCB traces carrying switching currents as short as possible to minimize current loops. Use ground planes for best results. Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Use a multilayer board whenever possible for better noise immunity and power dissipation. Follow these guidelines for good PCB layout: * Use a large copper plane under the MAX16812 package. Ensure that all heat-dissipating components have adequate cooling. Connect the exposed pad of the device to the ground plane. * Isolate the power components and high-current paths from sensitive analog circuitry. * Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. Keep switching loops short. * Connect AGND and SGND to a ground plane. Ensure a low-impedance connection between all ground points. * Keep the power traces and load connections short. This practice is essential for high efficiency. Use thick copper PCBs to enhance full-load efficiency. * Ensure that the feedback connection to FB is short and direct. * Route high-speed switching nodes away from the sensitive analog areas. * To prevent discharge of the compensation capacitors, CCOMP1 and CCOMP2, during the off-time of the dimming cycle, ensure that the PCB area close to these components has extremely low leakage.
18
______________________________________________________________________________________
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
Typical Application Circuit
BUCK-BOOST CONFIGURATION CH_REG
MAX16812
RCS
DOUT
VOUT
CSVIN LV IN CIN1 RT CL_REG L_REG RTGRM TGRM CTGRM DIM OV VOUT ROV1 ROV2 EN RT
CS+
DGT
DD
H_REG
HV
LX SRC GT
RSRC
COUT
RG DRV
MAX16812
SLP CSLP
SGND AGND
REF
REFI
CS_OUT
COMP FB
CREF
RCOMP1 RREF1 RCOMP2
CCOMP1
RREF2 CCOMP2
Pin Configuration
DRV GT HV
Chip Information
PROCESS: BiCMOS TRANSISTOR COUNT: 8699
TOP VIEW
LX LX 21 SRC 22 SRC 23 SLP 24 TGRM 25 DIM 26 RT 27 OV 28 1 FB 2 COMP 3 REFI 20 19
H_REG 16
18
17
15 14 13 12 CSCS+ DGT DD SGND L_REG IN
LV 11 10 9 8 7 EN
MAX16812
+
*EP
4 REF
5 CS_OUT
6 AGND
*EP = EXPOSED PAD
TQFN
______________________________________________________________________________________
19
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control MAX16812
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
PACKAGE OUTLINE, 16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm
21-0140
K
1
2
20
______________________________________________________________________________________
QFN THIN.EPS
Integrated High-Voltage LED Driver with Analog and PWM Dimming Control
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
MAX16812
PACKAGE OUTLINE, 16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm
21-0140
K
2
2
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 21
(c) 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.
Heaney

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